This invention relates to a switching power supply circuit which is provided as a power supply in various pieces of electronic equipment.
Meanwhile, the assignee of the present patent application has proposed various power supply circuits wherein a resonance type converter is provided on the primary side.
FIG. 14 is a circuit diagram showing an example of a switching power supply circuit which includes a resonance type converter and is formed based on the invention applied for patent in advance by the assignee of the present application.
Referring to FIG. 14, the power supply circuit shown includes a switching converter configured such that a partial voltage resonance circuit which performs a voltage resonance operation only upon turning off during switching is combined with a separately excited current resonance converter in which a half bridge coupling method is used.
First, in the power supply circuit shown in FIG. 14, a common mode noise filter formed from two filter capacitors CL and a common mode choke coil CMC is connected to a commercial AC power supply AC.
Further, as a rectification smoothing circuit for producing a DC input voltage from the commercial AC power supply AC, a full wave rectification circuit formed from a bridge rectification circuit Di and a smoothing capacitor Ci is provided at the following stage of the common mode noise filter.
A rectification output of the bridge rectification circuit Di is charged into the smoothing capacitor Ci. Consequently, across the smoothing capacitor Ci, a rectified smoothed voltage Ei (DC input voltage) equal to the AC input voltage VAC is obtained.
The current resonance capacitor which receives the DC input voltage as an input thereto to perform switching includes a switching circuit system formed from two switching devices Q1 and Q2 each in the form of a MOS-FET which are coupled by half bridge coupling as seen in FIG. 14. Damper diodes DD1 and DD2 formed from body diodes are connected in parallel in a direction shown in FIG. 14 between the drain-source of the switching devices Q1 and Q2, respectively.
A partial resonance capacitor Cp is connected in parallel between the drain and the source of the switching device Q2. A parallel resonance circuit (partial voltage resonance circuit) is formed from the capacitance of the partial resonance capacitor Cp and the leakage inductance L1 of a primary winding N1. By the partial voltage resonance circuit, a partial voltage resonance operation wherein voltage resonance occurs only upon turning off of the switching devices Q1 and Q2 is obtained.
In the power supply circuit, in order to switching drive the switching devices Q1 and Q2, an oscillation and driving circuit 2 formed from, for example, a general purpose IC is provided. The oscillation and driving circuit 2 includes an oscillation circuit and a driving circuit, and applies a drive signal (gate voltage) having a predetermined frequency to the gates of the switching devices Q1 and Q2. Consequently, the switching devices Q1 and Q2 perform the switching operation so as to alternately change over between an on state and an off state in the predetermined switching frequency.
An insulating converter transformer PIT (Power Isolation Transformer) transmits the switching output of the switching devices Q1 and Q2 to the secondary side.
A primary winding N1 of the insulating converter transformer PIT is connected at an end thereof to a node (switching output point) between the source of the switching device Q1 and the drain of the switching device Q2 through a primary side series resonance capacitor C1. Consequently, a switching output can be obtained.
The primary winding N1 is connected at the other end thereof to the primary side ground as shown in FIG. 14.
The series resonance capacitor C1 and the primary winding N1 are connected in series to each other. In particular, a primary side series resonance circuit for making the operation of the switching converter that of the current resonance type is formed from the capacitance of the series resonance capacitor C1 and the leakage inductance L1 of the primary winding N1 (series resonance winding) of the insulating converter transformer PIT.
According to the forgoing description, the primary side switching converter shown in FIG. 14 performs an operation as that of the current resonance type by the primary side series resonance circuit (L1-C1) and a partial voltage resonance operation by the partial voltage resonance circuit (Cp//L1) described above.
In particular, the power supply circuit shown in FIG. 14 is formed such that a resonance circuit for making a primary side switching converter as that of the resonance type and a different resonance circuit are combined with each other. Here, such a switching converter as just described is hereinafter referred to as composite resonance type converter.
While description given with reference to the drawings is omitted, the insulating converter transformer PIT includes an EE type core formed by combining E type cores formed from, for example, a ferrite material. A winding portion of the EE type core is divided into portions of the primary side and the secondary side, and the primary winding N1 and a secondary winding N2 are wound around an inner magnetic leg of the EE type core.
Further, a gap of 1.0 mm or less is formed in the inner magnetic leg of the EE type core of the insulating converter transformer PIT such that a coupling coefficient of 0.80 or more is obtained between the primary and secondary windings N1 and N2.
Actually, the gap G is set to G=1.0 mm, and the turn number of the primary winding N1 is set to N1=35 T (turns) and the turn number of the secondary winding N2 is set to N2=8 T so that a coupling coefficient K=approximately 0.80 can be obtained.
A center tap is provided for the secondary winding N2 of the insulating converter transformer PIT and connected to the secondary side ground as shown in FIG. 14. Further, a full wave rectification circuit is provided for the secondary winding N2 of the insulating converter transformer PIT and is formed from a rectification diode Do1, another rectification diode Do2 and a smoothing capacitor Co.
Consequently, as a voltage across the smoothing capacitor Co, a secondary side DC output voltage Eo which is a DC voltage of a level equal to the AC voltage excited by the secondary winding N2 is obtained. The secondary side DC output voltage Eo is supplied as a main DC power supply to a main load not shown, and is branched and inputted also as a detection voltage for constant voltage control to a control circuit 1.
It is to be noted that, in this instance, the rectification diodes Do1 and Do2 which form a full wave rectification circuit are actually formed from a single element in the form of a twin Schottky barrier diode TSD as indicated by a broken line framework in FIG. 14.
The control circuit 1 outputs a control signal as a voltage or current whose level is adjusted in response to the level of the secondary side DC output voltage Eo to the oscillation and driving circuit 2.
In the oscillation and driving circuit 2, the frequency of a switching driving signal to be applied to the gates of the switching devices Q1 and Q2 is varied based on the control signal inputted from the control circuit 1 so that an oscillation signal frequency produced by the oscillation circuit in the oscillation and driving circuit 2 is adjusted. Consequently, the switching frequency is adjusted. In this manner, since the switching frequency of the switching devices Q1 and Q2 is adjustably controlled in response to the level of the secondary side DC output voltage Eo, also the resonance impedance of the primary side DC resonance circuit is varied and the energy to be transmitted from the primary winding N1 which forms the primary side series resonance circuit to the secondary side is adjusted. Therefore, also the level of the secondary side DC output voltage Eo is adjustably controlled. As a result, constant voltage control of the secondary side DC output voltage Eo is implemented.
It is to be noted that the constant voltage controlling method of adjustably controlling the switching frequency to achieve stabilization in this manner is hereinafter referred to as “switching frequency controlling method”.
FIG. 15 is a waveform diagram illustrating operation of several components of the power supply circuit shown in FIG. 14. In FIG. 15, the waveforms on the left side indicate operation at the load power Po=150 W, but the waveforms on the right side indicate operation at the load power Po=25 W. The input voltage condition is set to the AC input voltage VAC=100 V fixed.
It is to be noted that, in this instance, the secondary side DC output voltage Eo is produced so as to have a voltage of 25 V.
Further, in the circuit shown in FIG. 14, in response to such a load condition and an input voltage condition as described above, several components are selectively set as follows:                insulating converter transformer PIT: gap G=1.0 mm, coupling coefficient k=0.80        primary winding N1=35 T        secondary winding N2=8 T (4 T+4 T across the center tap)        primary side series resonance capacitor C1=0.047 μF        partial resonance capacitor Cp=330 pF        
First, a voltage V1 indicated by a rectangular waveform in FIG. 15 is a voltage across the switching device Q2, and indicates on/off timings of the switching device Q2.
A period of time within which the level of the voltage V1 is 0 is an on period within which the switching device Q2 conducts. Within the on period, switching current IQ2 shown by the waveform in FIG. 15 flows to the switching circuit system formed from the switching device Q2 and the clamp diode DD2. Further, a period of time within which the voltage V1 is clamped to the level of the rectified smoothed voltage Ei is a period of time within which the switching device Q2 is off, and the level of the switching current IQ2 is 0 as seen in FIG. 15.
Further, though not shown in the drawings, a voltage across the switching device Q1 and switching current to be supplied to the switching circuit (Q1, DD1) are obtained with waveforms wherein the phases thereof are shifted by 180 degrees from those of the voltage V1 and the switching current IQ2. In particular, as described above, the switching devices Q1 and Q2 perform a switching operation at timings at which they are changed over between on and off alternately.
Further, primary side series resonance current Io to flow to the primary side series resonance circuit (C1-N1(L1)) is produced by combining the switching current flowing in the switching circuit (Q1, DD1) and the switching current flowing in the switching circuit (Q2, DD2), and the resulting current flows in accordance with the waveform shown in FIG. 15.
Further, for example, if the waveforms of the voltage V1 shown in FIG. 15 when the load power Po=150 W and when the load power Po=25 W are compared with each other, then it is recognized that the switching frequency on the primary side when the secondary side DC output voltage Eo is in a heavy load condition (Po=150 W) is controlled so as to be lower than that when the secondary side DC output voltage Eo is in a light load condition (Po=25 W). In particular, the switching frequency is controlled so as to become low in response to a drop of the level of the secondary side DC output voltage Eo when a heavy load condition is entered, but become high in response to an increase of the level of the secondary side DC output voltage Eo when a light load condition is entered. This indicates the fact that the constant voltage controlling operation by upper side control is performed as a switching frequency controlling method.
Further, by performing the operation on the primary side described above, an AC voltage V2 having a waveform shown in FIG. 15 is induced in the secondary winding N2 of the insulating converter transformer PIT. Then, within a period of one of half cycles within which the waveform of the AC voltage V2 indicates the positive polarity, the rectification diode Do1 on the secondary side conducts to allow rectification current ID1 to flow with the waveform and at the timing shown in FIG. 15. Further, within a period of the other half cycle within which the waveform of the AC voltage V2 indicates the negative polarity, the rectification diode Do2 on the secondary side conducts to allow rectification current ID2 to flow with the waveform and at the timing shown in FIG. 15. Further, in the full wave rectification circuit on the secondary side, rectification output current I2 flowing between the center tap of the secondary winding N2 and the secondary side ground is produced by combining the rectification current ID1 and the rectification current ID2 as seen in FIG. 15.
FIG. 16 is a graph illustrating the AC to DC power conversion efficiency with respect to the load variation and a characteristic of the switching frequency of the power supply circuit shown in FIG. 14 under the input voltage condition of the AC input voltage VAC=100 V.
First, the switching frequency fs decreases as the load becomes heavier in response to performance of the constant voltage controlling operation. However, this is not a characteristic that the switching frequency fs linearly varies with respect to the load variation, but the switching frequency fs is inclined to increase steeply within a range, for example, from the load power Po=approximately 25 W to Po=0 W.
Further, the AC→DC power conversion efficiency (ηAC→DC) is inclined to increase as the load power Po increases, and particularly when the load power Po=150 W, a result is obtained that AC to DC power conversion efficiency ηAC→DC is 90% or more.
Incidentally, where the configuration as a resonance type converter which stabilizes the secondary side DC output voltage by the switching frequency controlling method is applied as in the power supply circuit shown in FIG. 14, the variable control range of the switching frequency for stabilization is inclined to be a comparatively wide range.
This is described with reference to FIG. 17. FIG. 17 illustrates the constant voltage control characteristic of the power supply circuit shown in FIG. 14 as a relationship between the switching frequency fs and the level of the secondary side DC output voltage Eo.
It is to be noted that the description given with reference to FIG. 17 presupposes that the upper side control is adopted as the switching frequency controlling method by the power supply circuit shown in FIG. 14. Here, the upper side control is a control method wherein the switching frequency is variably controlled within a frequency range higher than the resonance frequency fo of the primary side series resonance circuit such that the level of the secondary side DC output voltage Eo is controlled making use of the variation of the resonance impedance caused by the variable control of the switching frequency.
Generally, a series resonance circuit exhibits the lowest resonance impedance at the resonance frequency fo. Consequently, as a relationship in the upper side control between the secondary side DC output voltage Eo and the switching frequency fs, the level of the secondary side DC output voltage Eo increases as the switching frequency fs comes nearer to the resonance frequency fo1, but decreases as the switching frequency fs goes away from the resonance frequency fo1.
Accordingly, as seen in FIG. 17, the level of the secondary side DC output voltage Eo with respect to the switching frequency fs under the condition that the load power Po is fixed indicates such a quadratic curve variation that the level exhibits a peak when the switching frequency fs is equal to the resonance frequency fo1 of the primary side series resonance circuit but decreases as the switching frequency fs goes away from the resonance frequency fo1.
Further, where the level of the secondary side DC output voltage Eo at the minimum load power Pomin and the level of the secondary side DC output voltage Eo at the maximum load power Pomax corresponding to the same switching frequency fs are compared with each other, a characteristic that the level of the secondary side DC output voltage Eo is shifted so as to decrease by a predetermined amount can be obtained at the maximum load power Pomax rather than at the minimum load power Pomin. In particular, where it is considered that the switching frequency fs is fixed, the level of the secondary side DC output voltage Eo decreases as the load becomes heavier.
Then, if it is attempted under such a characteristic as described above to stabilize the secondary side DC output voltage Eo so as to be Eo=tg by the upper side control, then the variable range (necessary control range) of the switching frequency necessary for the power supply circuit shown in FIG. 14 is a range indicated by reference character Δfs in FIG. 17.
Actually, the power supply circuit shown in FIG. 14 performs constant voltage control so that the secondary side DC output voltage Eo may be stabilized, for example, at the secondary side DC output voltage Eo=25 V using the switching frequency controlling method in accordance with the input variation range of the AC input voltage VAC=85 V to 120 V of the AC 100 V system and the load conditions of the maximum load power Pomax=150 W and minimum load power Pomin=0 W (no load) to the secondary side DC output voltage Eo which is the main DC power supply.
In this instance, the variable range of the switching frequency fs varied by the power supply circuit shown in FIG. 14 in order to perform constant voltage control is a range from fs=80 kHz to 200 kHz or more, and also the variable range Δfs is 120 kHz or more and is a wide range in its own way.
A power supply circuit formed so as to be capable of operating in response to, for example, an AC input voltage range of approximately AC 85 V to 288 V so that the power supply circuit can be ready for, for example, areas of the AC input voltage AC 100 V system such as Japan, U.S.A and so forth and areas of the AC 200 V system such as Europe and so forth, that is, a power supply circuit ready for a wide range, is known.
Thus, it is examined to form the power supply circuit shown in FIG. 14 as a power supply circuit ready for the wide range described above.
The power supply circuit ready for the wide range is ready for, for example, the AC input voltage range of AC 85 V to 288 V as described above. Accordingly, when compared with an alternative case wherein the power supply circuit is ready for a single range of, for example, only the AC 100 V system or only the AC 200 V system, also the variation range of the level of the secondary side DC output voltage Eo becomes great. In order to carry out constant voltage control for such a secondary side DC output voltage Eo having a level variation range increased corresponding to the wide AC input voltage range as just described, an increased switching frequency control range is required. For example, in the circuit shown in FIG. 14, it is necessary to expand the control range of the switching frequency fs to a range of approximately 80 kHz to 500 kHz.
However, in an existing IC (oscillation and driving circuit 2) for driving a switching device, the upper limit to the driving frequency for which it is ready is approximately 200 kHz. Further, even if a switching driving IC which can drive at such a high frequency as described above is formed and mounted, where a switching device is driven at such a high frequency as described above, the power conversion efficiency decreases remarkably. Therefore, the switching driving IC cannot be practically used as an actual power supply circuit. It is to be noted that the upper limit to the level of the AC input voltage VAC which can be stabilized, for example, by the power supply circuit shown in FIG. 14 is approximately 100 V.
Therefore, it is known that, if it is tried to make a switching power supply circuit, which uses the switching frequency control method for stabilization, actually ready for the wide range, then, for example, such countermeasures as described just below are taken.
As one of the countermeasures, a rectification circuit system for receiving a commercial AC power supply as an input thereto to produce a DC input voltage (Ei) is provided with a function of performing changeover between a voltage doubler rectification circuit and a full wave rectification circuit in response to an input of the commercial AC power supply of the AC 100 V system or the AC 200 V system.
In this instance, the circuit is formed such that the commercial AC power supply level is detected and the circuit connection in the rectification circuit system is changed over in response to the detected level by a switch in which electromagnetic relays are used so as to form the voltage doubler rectification circuit or the full wave rectification circuit.
However, in such a configuration as just described which involves changeover of the rectification circuit, a required number of electromagnetic relays are required as described above. Further, it is necessary to provide at least two smoothing capacitors in order to form the voltage doubler rectification circuit. Therefore, the cost is increased by increase of the number of components and also the mounting area of a circuit board of the power supply circuit is increased, which increases the scale of the power supply circuit. Particularly, since the smoothing capacitors and the electromagnetic relays are large among various components for forming the power supply circuit, the size of the circuit board becomes considerably great.
It is assumed here that, where the configuration wherein a full wave rectification operation and a voltage doubler rectification operation are changed over is used, if the level of the AC input voltage while a commercial AC power supply of the AC 200 V system is inputted becomes lower than that ready for the AC 200 V system because instantaneous interruption occurs or because the AC input voltage drops to a level lower than the rated voltage, then a malfunction occurs that changeover to the voltage doubler rectification circuit is performed because it is detected in error that an AC input voltage of the AC 100 V system is inputted. If such a malfunction as just described occurs, then voltage doubler rectification is performed for the AC input voltage actually of the level of the AC 200 V system. Therefore, the resulting voltage exceeds the withstanding voltage, for example, of the switching devices Q1 and Q2, and as a result, there is the possibility that the switching devices Q1 and Q2 may be broken.
Therefore, in order to prevent occurrence of such a malfunction as described above, an actual circuit is configured such that not only the DC input voltage of a switching converter which is a main switching converter but also the DC input voltage of a converter circuit on the standby power supply side are detected. Consequently, components for detecting the converter circuit on the standby power supply circuit side are additionally provided, and as a result, increase of the cost and increase of the size of the circuit board described above are further promoted.
Further, that the DC input voltage of the converter on the standby power supply side is detected in order to prevent the malfunction signifies that the power supply circuit which includes a circuit for changing over the rectification operation and is ready for the wide range can be used actually only for electronic equipment which includes not only a main power supply but also a standby power supply. In particular, the type of an electronic apparatus capable of incorporating the power supply is limited to that of an electronic apparatus which includes a standby power supply, and as a result, the utilization range decreases as much.
Further, as one of configurations ready for the wide range, also a configuration is known wherein the type of the current resonance converter on the primary side is changed over between the half bridge coupling type and the full bridge coupling type in response to an input of a commercial AC power supply of the AC 100 V system/AC 200 V system.
In the configuration described, even if the level of the AC input voltage of the AC 200 V system decreases to that of the AC 100 V system, for example, as a result of instantaneous interruption or the like as described above and causes a malfunction, only the switching operation is changed over from the half bridge operation to the full bridge operation, but the withstanding voltage of the switching device and so forth is not exceeded. Therefore, the DC input voltage on the standby power supply side need not be detected, and the present configuration can be applied to an electronic apparatus which does not include a standby power supply. Further, since the changeover in the configuration is not that on the commercial power supply line and the circuit form can be changed over by a semiconductor switch, a large-size switching member such as an electromagnetic relay need not be provided.
However, according to the configuration described above, in order to form the full bridge coupling so as to implement the configuration ready for an input of the AC 100 V system, it is necessary to provide at least four switching devices. In other words, when compared with the configuration of a converter which can be formed from two switching devices and to which only the half bridge coupling method is applied, two additional switching devices must be provided.
Further, according to the configuration, four switching devices perform the switching operation in the full bridge operation, but also in the half bridge operation, three switching devices perform the switching operation. While the resonance converter can be operated with low switching noise, as the number of switching devices which perform switching in this manner increases, the disadvantage increases as regards switching noise.
Also where any one of the configurations described above is adopted as a configuration ready for the wide range, when compared with a configuration ready for a single range, increase of the cost and increase of the circuit size arising from increase of the number of parts or the like cannot be avoided. Further, new problems which do not appear with the configuration ready for a single range such as a limit to the utilization range to equipment in the case of the former configuration and increase of switching noise in the case of the latter configuration and so forth appear.
Further, where the control range of the switching frequency is suitably wide as in the power supply circuit shown in FIG. 14, also a problem appears that the high speed response characteristic of stabilization of the secondary side DC output voltage Eo degrades.
Depending upon an electronic apparatus, there is the possibility that the load condition may vary in such a manner as to instantaneously change over, for example, between a maximum load state and a substantial no load state. A load which exhibits such a load variation as just described is sometimes called switching load. The power supply circuit to be incorporated in an apparatus which serves as a switching load as just described must be configured so that the secondary side DC output voltage is optimally stabilized so that it is ready also for the load variation of the switching load described above.
However, where the control range of the switching frequency is wide as described with reference to FIG. 17, in order to adjust the switching frequency to a switching frequency with which the secondary side DC output voltage is adjusted to a required level in response to the load variation of a load like such a switching load as described above, a comparatively long period of time is required. In short, an unfavorable result is obtained as the response characteristic of the constant voltage control.
Particularly, as shown in FIG. 16, as the switching frequency characteristic by constant voltage control by the power supply circuit shown in FIG. 14, the switching frequency varies by a great amount within the load range of the load power Po from Po=approximately 25 W to Po=0 W. Therefore, it is recognized that the power supply circuit is disadvantageous in the responsibility in constant voltage control for such a switching load as described above.
It is desirable to provide a power supply circuit which performs constant voltage control by switching frequency control and is ready for a wide range while the necessary control range of the switching frequency control is reduced.